電流型測(cè)試儀表放大器提升壓電加速度計(jì)性能

電路抑制由壓電傳感器及其電纜產(chǎn)生的容性耦合噪聲。

一個(gè)典型的壓電傳感器由表面上金屬化電極的PZT-5A陶瓷材料組成。在電極處使用導(dǎo)電環(huán)氧將傳感器連接到外部電纜。絕緣膠粘合裝置元件到待測(cè)結(jié)構(gòu)上，使傳感器與接地參考電位隔離。壓電片面向預(yù)期加速度的方向。當(dāng)安放在目標(biāo)結(jié)構(gòu)上時(shí)，壓電片成為簡(jiǎn)單的壓力傳感器和加速度計(jì)，產(chǎn)生正比于壓力且平行于壓電片極化方向的電壓。壓電片容性阻抗在低頻時(shí)呈現(xiàn)很大的電抗，使壓電片和電纜易受周圍電氣設(shè)備和電源線的干擾。傳感器遠(yuǎn)距離安放時(shí)，需要使用屏蔽的互連電纜，但即使屏蔽，對(duì)去除共模信號(hào)也不是完全有效，因?yàn)閴弘娖膶?dǎo)電表面仍會(huì)獲取噪聲。

提取傳感器信號(hào)的一個(gè)方法是使用儀表放大器，它只放大傳感器所產(chǎn)生的電位。放大器抑制出現(xiàn)在傳感器各端的共模耦合噪聲電位。

典型的微型壓電片傳感器直徑0.125英寸，0.0075英寸厚，相當(dāng)于幾乎500 pF的電容。如果測(cè)量應(yīng)用需要限制激勵(lì)頻率在10Hz或更低的動(dòng)態(tài)響應(yīng)，傳感器輸出電抗可達(dá)到10幾MΩ的范圍。電路印制板的絕緣層和周圍濕度使放大器輸入阻抗有幾乎10 MΩ的實(shí)際限制。

必須謹(jǐn)慎選擇絕緣方法和使用保護(hù)電位，必須使用輸入偏置電流為微微安級(jí)的放大器。否則，傳感器電容和放大器的輸入偏置電流電阻，對(duì)儀表放大器的信號(hào)施加相位偏移。為消除保護(hù)和復(fù)雜的絕緣需要，圖1電路使用帶反饋的儀表放大器，測(cè)量傳感器短路電流，而不是開路電壓。傳感器和信號(hào)地之間的共模電壓VCM來自周圍雜散電容耦合帶來的噪聲源。下面的公式描述了傳感器輸出電流i和其開路輸出電壓ES的關(guān)系：

圖1中，A代表IC1的電壓增益，R=R1=R2。電阻R1和R2為IC1(INA121儀表放大器)提供反饋和輸入偏置電流回路，電阻RG設(shè)置放大器增益。INA121的輸入偏置電流為0.5pA，在10MΩ反饋電阻上產(chǎn)生5µV電壓偏置。放大器增益為500倍時(shí)，IC1輸出偏置達(dá)到2.5 mV。放大器IC2 TL081提供單位增益的信號(hào)極性變換。

如果2A+1>>2RjωCS，則i≈jωCSES。放大器IC1輸入電壓V1變?yōu)榱?，因?yàn)榉糯笃鬏斎虢K端通過傳感器起虛短電路的作用。取儀表放大器和反相放大器輸出、兩個(gè)反饋電阻和儀表放大器輸入端子構(gòu)成回路的電壓和(其電位差為零)，得到eO=jωRCES，其中eO表示IC1的輸出，也是IC2輸出的負(fù)值。

在下面的公式中，用運(yùn)算放大器做的積分器IC3給出在IC3輸出的ES。

圖1中的元器件值，IC1提供500倍增益。電阻R1和R2為10 MΩ，壓電傳感器的電容為500 pF。對(duì)最高頻率10Hz，量值為2RωCS=0.62A+1=501和傳感器輸出ES，也沒有相位誤差E'。電路能測(cè)量準(zhǔn)靜態(tài)壓力的改變;電路能維持C1上的電荷，從而對(duì)電路頻率響應(yīng)構(gòu)成限制。

英文原文：

Circuit suppresses capacitively coupled noise pickup by piezoelectric sensor and its wiring.

Dave Wuchinich, Modal Mechanics, Yonkers, NY; Edited by Brad Thompson and Fran Granville -- EDN, 11/23/2006

A typical piezoelectric sensor comprises a disk of PZT-5A ceramic material with metallized electrodes on its

surfaces. Applying electrically conductive epoxy to the electrodes connects external wiring to the sensor. An insulating adhesive attaches the assembly to the structure under test and isolates the sensor from ground-referenced potentials. The disk faces the direction of the expected acceleration. When you mount the piezoelectric disk on a target structure, it serves as a simple force sensor and accelerometer by producing a voltage that's directly proportional to the force acting parallel to the disk's direction of polarization. A piezoelectric disk's capacitive impedance presents a large reactance at low frequencies, making the disk and its wiring susceptible to interference that surrounding electrical equipment and power lines produce. Placing the sensor in a remote location requires shielded interconnecting cable, but even shielding is not entirely effective in removing common-mode signals because noise pickup can still occur at the disk's conductive surfaces.

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One method of extracting the sensor's signal employs an instrumentation amplifier, which amplifies only the potential the sensor produces; the amplifier rejects common-mode-coupled noise potential that appears on each of the sensor's terminals.

A typical miniature piezoelectric- disk sensor that's 0.125 in. in diameter and 0.0075 in. thick presents a capacitance of approximately 500 pF. If the measurement application requires a dynamic response to force excitation frequencies of 10 Hz or below, the sensor's output reactance ranges into the tens of megohms. The circuit's pc-board insulating substrate and ambient humidity impose a practical limit of approximately 10 MΩ on the amplifier's input resistance.

You must carefully choose insulation and apply guarding potentials, and you must use an amplifier with picoampere input-bias currents. Otherwise, the sensor's capacitance and the amplifier's input-bias-current resistors impose a phase shift on the signal you apply to the instrumentation amplifier. To eliminate guarding and elaborate insulation requirements, the circuit in Figure 1 uses an instrumentation amplifier with feedback to measure the sensor's short-circuit current and not its open-circuit voltage. VCM, the common-mode voltage between the sensor and the signal ground, results from nearby noise sources resulting from stray capacitive coupling. The following equation relates the sensor's output current, i, and its open-circuit output voltage, ES:

where A represents IC1's voltage gain, and R=R1=R2 in Figure 1. Resistors R1 and R2 provide feedback and input-bias-current-return paths for IC1, an INA121 instrumentation amplifier, and resistor RG sets the amplifier's gain. The INA121's input-bias-offset current of 0.5 pA produces 5 µV of voltage offset across its 10-MΩ feedback resistors. At an amplifier gain of 500, IC1's output offset amounts to 2.5 mV. Amplifier IC2, a TL081, provides unity-gain signal inversion.

If 2A+1>>2R

jωCS, then i≈jωCSES, and amplifier IC1's input voltage, VI, vanishes because the amplifier's input terminals act as a virtual short circuit across the sensor. Taking the sum of voltages around the loop comprising the instrumentation and inverting amplifiers' output, the two feedback resistors and the instrumentation amplifier's input terminals, whose potential difference is zero, yields eO=jωRCES, where eO represents IC1's output and also the negative value of IC2's output.

An operational-amplifier-based integrator, IC3, delivers the value for ES at IC3's output, E' in the following equation.

For the component values in Figure 1, IC1 provides a gain of 500. Resistors R1 and R2 are equal at 10 MΩ, and the piezoelectric sensor's capacitance measures 500 pF. For the highest frequency of interest, 10 Hz, the quantity 2RωCS=0.62A+1=501 and the sensor's output, ES, appear without phase error as E'. This circuit can measure quasistatic force changes; the circuit's ability to sustain a charge on C1 imposes the ultimate limit on the circuit's frequency response.